LT3799EMSE-1-PBF Просмотр технического описания (PDF) - Linear Technology
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LT3799EMSE-1-PBF Datasheet PDF : 20 Pages
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LT3799-1
OPERATION
Table 1. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted
TRANSFORMER SIZE
PART NUMBER (L × W × H)
JA4429
21.1mm × 21.1mm × 17.3mm
7508110210 15.75mm × 15mm × 18.5mm
750813002
15.75mm × 15mm × 18.5mm
750811330
43.2mm × 39.6mm × 30.5mm
750813144
16.5mm × 18mm × 18mm
750813134
16.5mm × 18mm × 18mm
750811291
31mm × 31mm × 25mm
750813390
43.18mm × 39.6mm × 30.48mm
750811290
31mm × 31mm × 25mm
X-11181-002 23.5mm × 21.4mm × 9.5mm
LPRI
NPSA
RPRI
(μH)
(NP:NS:NA)
(mΩ)
400
1:0.24:0.24
252
2000
6.67:1:1.67
5100
2000
20:1.0:5.0
6100
300
6:1.0:1.0
150
600
4:1:0.71
2400
600
8:1:1.28
1850
400
1:1:0.24
550
100
1:1:0.22
150
460
1:1:0.17
600
500
72:16:10
1000
RSEC
(mΩ)
126
165
25
25
420
105
1230
688
560
80
MANUFACTURER
Coilcraft
Würth Elektronik
Würth Elektronik
Würth Elektronik
Würth Elektronik
Würth Elektronik
Würth Elektronik
Würth Elektronik
Würth Elektronik
Premo
TARGET
APPLICATION
(VOUT / IOUT)
22V/1A
10V/0.4A
3.8V/1.1A
18V/5A
28V/0.5A
14V/1A
85V/0.4A
90V/1A
125V/0.32A
30V/0.5A
Loop Compensation
The current output feedback loop is an integrator con-
figuration with the compensation capacitor between the
negative input and output of the operational amplifier.
This is a one-pole system therefore a zero is not needed
in the compensation. For offline applications with PFC,
the crossover should be set an order of magnitude lower
than the line frequency of 120Hz or 100Hz. In a typical
application, the compensation capacitor is 0.1μF.
In non-PFC applications, the crossover frequency may
be increased to improve transient performance. The
desired crossover frequency needs to be set an order
of magnitude below the switching frequency for optimal
performance.
MOSFET and Diode Selection
With a strong 1.9A gate driver, the LT3799-1 can effectively
drive most high voltage MOSFETs. A low Qg MOSFET is
recommended to maximize efficiency. In most applications,
the RDS(ON) should be chosen to limit the temperature rise
of the MOSFET. The drain of the MOSFET is stressed to
VOUT • NPS + VIN during the time the MOSFET is off and
the secondary diode is conducting current. But in most
applications, the leakage inductance voltage spike exceeds
this voltage. The voltage of this stress is determined
by the switch voltage clamp. Always check the switch
waveform with an oscilloscope to make sure the leakage
inductance voltage spike is below the breakdown voltage
of the MOSFET. A transient voltage suppressor and diode
are slower than the leakage inductance voltage spike,
therefore causing a higher voltage than calculated.
The secondary diode stress may be as much as
VOUT + 2 • VIN/NPS due to the anode of the diode ringing
with the secondary leakage inductance. An RC snubber
in parallel with the diode eliminates this ringing, so that
the reverse voltage stress is limited to VOUT + VIN/NPS.
With a high NPS and output current greater than 3A, the
IRMS through the diode can become very high and a low
forward drop Schottky is recommended.
37991f
15
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